Wireless-transmitter circuits including power digital-to-amplitude converters

ABSTRACT

Circuits comprising: digital-to-amplitude converter (DAC), comprising: binary weighted switching transistors (BWSTs), each having gate coupled to amplitude control bit ACB, and wherein the drain of each of the BWSTs are connected together and wherein the source of each of the BWSTs are connected together; transistor M 1  having gate coupled to input signal and first bias voltage BV 1  and source coupled to the drains of the BWSTs; transistor M 2  having gate coupled to BV 2  and source coupled to the drain of M 1;  transistor M 3  having gate coupled to BV 3  and source coupled to the drain of M 2;  transistor having gate coupled to BV 4,  source coupled to the drain of M 3;  and inverter having input coupled to another ACB and having output coupled to the output of the DAC and the drain of M 4.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No. 14/873,177, filed Oct. 1, 2015, which claims the benefit of United States Provisional Patent Application No. 62/058,603, filed Oct. 1, 2014, each of which is hereby incorporated by reference herein in its entirety.

STATEMENT REGARDING GOVERNMENT FUNDED RESEARCH

This invention was made with government support under contract FA8650-10-1-7042 awarded by Defense Advanced Research Projects Agency. The government has certain rights in the invention.

BACKGROUND

As the number of electronic devices wirelessly communicating with devices connected to the Internet and each other continues to increase, the need to improve mechanisms for long-range, high-data-rate wireless communication similarly increases. Areas for improvement of existing technologies include, for example, the cost of transmitters and receivers, the power used, the range of communications, the size of the transmitters and receivers, the ability to reduce interference between transmitters and receivers, etc.

Accordingly, new circuits and methods for wireless transmitters are provided.

SUMMARY

Circuits and methods for wireless transmitters are provided. In some embodiments, circuits for a transmitter are provided, the circuits comprising: a digital-to-amplitude converter (DAC) having an input and an output, comprising: a plurality of binary weighted switching transistors, each having a gate coupled to one of a plurality of amplitude control bits, a drain, and a source, and wherein the drain of each of the plurality of binary weighted switching transistors are connected together and wherein the source of each of the plurality of binary weighted switching transistors are connected together; a first transistor having a gate coupled to an input signal and a first bias voltage, a source coupled to the drains of the plurality of binary weighted switching transistors, and a drain; a second transistor having a gate coupled to a second bias voltage, a source coupled to the drain of the first transistor, and a drain; a third transistor having a gate coupled to a third bias voltage, a source coupled to the drain of the second transistor, and a drain; a fourth transistor having a gate coupled to a fourth bias voltage, a source coupled to the drain of the third transistor, and a drain; and an inverter having an input coupled to another amplitude control bit and having an output coupled to the output of the DAC and the drain of the fourth transistor; and an antenna coupled to the output of the DAC.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of an example of a transmitter in accordance with some embodiments.

FIG. 2 is a schematic of an example of a frequency multiplier in accordance with some embodiments.

FIG. 3 is a schematic of an example of phase modulator/shifter in accordance with some embodiments.

FIG. 4 is a schematic of an example of a mixer in accordance with some embodiments.

FIG. 5 is a schematic of an example of an array driver in accordance with some embodiments.

FIG. 6 is a schematic of an example of a limiting amplifier in accordance with some embodiments.

FIG. 7 is a schematic of an example of a hybrid power digital-to-amplitude converter in accordance with some embodiments.

FIG. 8 is a schematic of an example of a two-stack driver in accordance with some embodiments.

FIG. 9 is a schematic of an example of an adaptive bias circuit in accordance with some embodiments.

FIG. 10 is a schematic of an example of a four-stack amplifier in accordance with some embodiments.

DETAILED DESCRIPTION

In accordance with some embodiments, circuits and methods for wireless transmitters are provided.

Turning to FIG. 1, an example 100 of a digital polar phased array transmitter in accordance with some embodiments is shown. In some embodiments, any suitable number of transmitters 100 can be used in a transmitter application. For example, in some embodiments, one transmitter (having four (for example) elements (described below)) can be used. In another embodiment, four transmitters (each having four (for example) elements (described below)) can be used, for example.

In some embodiments, any suitable modulation technique can be used with transmitter(s) 100. For example, in some embodiments, QAM64 modulation can be used.

As illustrated, transmitter 100 includes a local oscillator reference input 102, a frequency multiplier 104, a quadrature hybrid 106, a resistor 108, a phase modulator 110, a digital interface 112, an array driver 114, digital polar transmitter elements 116, 118, 120, and 122, transmitter outputs 124, 126, 128, and 130, serial digital inputs 132, a global biasing circuit 170, and an ESD circuit 172.

A local oscillator reference signal is received by transmitter 100 at input 102. Any suitable local oscillator reference signal having any suitable frequency can be used. For example, in some embodiments, local oscillator reference signal can have a frequency of 30 GHz.

The local oscillator reference signal is received by frequency multiplier 104 and multiplied to a higher frequency. Any suitable frequency multiplier can be used (e.g., a frequency multiplier as described below in connection with FIG. 2 can be used), and the local oscillator reference signal can be multiplied by any suitable amount. For example, in some embodiments, the frequency multiplier can multiply the local oscillator reference signal by two.

The output of frequency multiplier 104 can be received by quadrature hybrid 106. The quadrature hybrid can be any suitable quadrature hybrid in accordance with some embodiments. As illustrated, resistor 108 can be connected from one of the inputs of the quadrature hybrid to ground to provide a reference impedance. Any suitable resistor can be used in some embodiments. For example, a 50 ohm resistor can be used.

In-phase and quadrature components of the multiplied local oscillator reference signal can be output by hybrid 106 to phase modulator 110. The phase modulator can be any suitable phase modulator, such as the phase modulator/shifter described below in connection with FIGS. 3 and 4. The phase modulator can be controlled by phase control outputs [P_(0,I)-P_(7,I)] and [P_(0,Q)-P_(7,Q)] of digital interface 112.

Array driver 114 can receive the output of phase modulator 110 and provide a drive signal to digital polar transmitter elements 116, 118, 120, and 122 that is split evenly among the digital polar transmitter elements. Any suitable array driver can be used in some embodiments. For example, in some embodiments, the array driver described below in connection with FIG. 5 can be used.

Digital polar transmitter elements 116, 118, 120, and 122 can drive transmitter outputs 124, 126, 128, and 130 in response to the drive signal from array driver 114 and amplitude control outputs [A₀-A₇] of digital interface 112. Each of transmitter outputs 124, 126, 128, and 130 can be connected to a suitable antenna. For example, in some embodiments, the antenna(s) can be phased array antennas, on-printed-circuit-board antennas, and/or any other suitable type of antenna. Any suitable number of digital polar transmitter elements can be used in some embodiments. For example, in some embodiments, four digital polar transmitter elements can be used to drive a 2×2 array of four antennas.

As further shown in FIG. 1, digital polar transmitter elements 116, 118, 120, and 122 can each include a resistor 140, a quadrature hybrid 142, a phase shifter 144, a limiting amplifier 146, and a hybrid power digital amplitude converter (DAC) 148.

The drive signal from array driver 114 can be provided to quadrature hybrid 142. The quadrature hybrid can be any suitable quadrature hybrid in accordance with some embodiments. As illustrated, resistor 140 can be connected from one of the inputs of the quadrature hybrid to ground to provide a reference impedance. Any suitable resistor can be used in some embodiments. For example, a 50 ohm resistor can be used.

In-phase and quadrature components of the drive signal from array driver 114 can be output by hybrid 142 to phase shifter 144. The phase shifter can be any suitable phase shifter, such as the phase modulator/shifter described below in connection with FIGS. 3 and 4. The phase shifter can be controlled by control signals φ₁, φ₂, φ₃, and φ₄ from a controller 174. These control signals can be used to control the phase of the signal to be transmitted by the digital polar transmitter element for any suitable purpose, such as for beamforming.

Controller can include any suitable hardware processor (e.g., a microprocessor, microcontroller, dedicated control logic, a digital signal processor, etc.), a scan chain, registers, memory, interfaces, inputs, outputs, etc. and can perform any suitable functions, such as controlling phase shifters 144, controlling bias functions, performing specialized processing for phased arrays, compensating for various implementation non-idealities that result in beam pointing error, etc.

The outputs of phase shifter 144 can be provided to limiting amplifier 146. The limiting amplifier can be any suitable limiting amplifier, such as the limiting amplifier described below in connection with FIG. 6.

The output of limiting amplifier 146 can be provided to hybrid power DAC 148. Hybrid power DAC 148 can be any suitable hybrid power DAC in some embodiments. For example, hybrid power DAC 148 can be implemented using the hybrid power DAC described below in connection with FIG. 7.

As shown in FIG. 1, digital interface 112 can include variable gain amplifiers (VGAs) 158, continuous time linear equalizers (CTLEs) 160, and demultiplexers 162, 164, and 166. Digital interface can receive digital serial inputs 132. More particularly, inputs 132 can include I phase control inputs, Q phase control inputs, amplitude control inputs, and a clock at inputs 150, 154, 152, and 156, respectively. Based on the inputs received at 132, the digital interface can generate phase control outputs [P_(0,I)-P_(7,I)] and [P_(0,Q)-P_(7,Q)] and amplitude control outputs [A₀-A₇] from demultiplexers 162, 166, and 164, respectively.

Global biasing circuitry 170 can be provided, as known in the art, to generate biasing voltages in circuit 100 in accordance with some embodiments.

ESD circuitry 172 can be provided, as known in the art, to protect circuit 100 from electrostatic discharge and over-voltage conditions in accordance with some embodiments.

FIG. 2 shows an example 200 of a frequency multiplier that can be used to implement frequency multiplier 104 of FIG. 1 in some embodiments. As shown, a local oscillator reference signal can be received at node 202, the signal multiplied by two, and then the resulting signal output at node 204. In some embodiments, match blocks 206 and 208 can include any suitable components for matching the impedance of the points on the left and right of each match block. For example, in some embodiments, match blocks 206 and 208 can include inductors, spirals, transmission lines, and/or capacitors.

Turning to FIG. 3, an example 300 of a phase modulator/shifter that can be used as phase modulator 110 and/or phase shifter 144 of FIG. 1 in some embodiments. As shown, phase modulator/shifter 300 includes mixers 302 and 304. Any suitable mixers can be used as mixers 302 and 304. For example, in some embodiments, example mixer 400 described below in connection with FIG. 4 can be used as mixers 302 and/or 304.

As shown in FIG. 4, mixer 400 includes eight switching transistors represented in this figure by transistors 402, 404, 406, and 408. The transistors other than transistor 408 (the most significant bit (MSB) transistor) are binary weighted with weights of W, 2W, . . . , 2 ⁶W, where W represents a given combination of finger width and number of fingers in a transistor. Any suitable combination of finger width and number of fingers can be used in the transistors for W, such as a finger width of 0.152 micron and one finger, in some embodiments.

The gates of these transistors are connected to inputs b₀, b₁, b₂, . . . , b₇. These bits can be provided by phase control bits [P_(0,I)-P_(7,I)], [P_(0,Q)-P_(7,Q)], φ1, φ2, φ3, φ4 shown in FIG. 1 in some embodiments. The binary value that is provided to the inputs, determines the amount of modulation or shift of the input signal such that a higher value turns on a higher total weighting of switches, resulting in a higher current flow through the switches. Although eight transistors and inputs are shown, any suitable number of inputs and transistors can be used in some embodiments. A bias voltage, V_(b), can also be provided to calibrate the shifter to account for variations in process, voltage, and temperature.

Turning to FIG. 5, an example 500 of an array driver that can be used to implement array driver 114 of FIG. 1 is shown in accordance with some embodiments. As shown, an input signal can be presented at inputs 502 of driver 500, the signal will be amplified, and the resulting signal will be provided at node 504.

Turning to FIG. 6, an example 600 of a limiting amplifier that can be used to implement limiting amplifier 146 of FIG. 1 is shown in accordance with some embodiments. As shown, an input signal can be presented at node 602 of limiting amplifier 600, the signal will be amplified, and the resulting signal will be provided at node 504.

FIG. 7 shows an example 700 of a hybrid power DAC that can be used to implement hybrid power DAC 148 of FIG. 1 in accordance with some embodiments. As shown, in some embodiments, hybrid power DAC 700 can be implemented as a differential Class-E power amplifier with four stacked transistors that is augmented with trail transistors (at the common source node) and a supply inverter (connected to the differential DC-feed spiral of the four-stack Class E power amplifier) to incorporate amplitude modulation capability.

In some embodiments, hybrid power DAC 700 includes inputs 702 and 704, outputs 706 and 708, inverters 710, switching transistors 712, 714, and 716, stacked transistors 717, 718, 720, and 722, DC feed inductors (implemented as transmission lines) 724, gate bias inputs 726, 728, 730, and 732, two-stack drivers 734, and match blocks 736.

As shown in FIG. 7, hybrid power DAC 700 includes two inverters 710 and N−1 switching transistors, represented in this figure by transistors 712, 714, and 716. The N−1 switching transistors (represented by transistors 712, 714, and 716) are binary weighted with weights of W₁, 2W₁, . . . , 2 ^(N−1)W₁, where W₁ represents a given combination of finger width and number of fingers in a transistor. Any suitable combination of finger width and number of fingers can be used in the transistors for W₁, such as a finger width of 2.793 micron and two fingers, in some embodiments. The gates of the transistors are connected to inputs b₀, b₁, b₂, . . . , b_(N−1) and the input to the inverter is connected to input b_(N) (the most significant bit (MSB)).

As suggested by the use of N in FIG. 7, any suitable number of switching transistors, and hence inputs b₀ . . . b_(N−1) can be used in some embodiments.

The binary value that is provided to inputs b₀, b₁, b₂, . . . , b_(N) determines the amount of amplification provided by the hybrid power DAC. These bits can be provided by amplitude control bits [A₀-A₇] shown in FIG. 1 in some embodiments. More particularly, a higher value at inputs b₀, b₁, b₂, . . . , b_(N−1) turns on a higher total weighting of the switching transistors, resulting in a higher current flow through the switching transistors, and the value of input b_(N) at the input to inverter 710 determines the supply voltage V_(DD,PA) supplied to the stack of transistors including stacked transistors 717, 718, 720, and 722 and the switching transistors.

The manner in which this amplification is achieved is further illustrated in connection with FIG. 10. As shown, for a given input at the gate of transistor M1 1002, a voltage between ground and 2V_(DD) (represented by voltage 1010) (where, V_(DD) here represents the nominal voltage supply in the technology used, for example, 1.2V nominally in 45 nm SOI CMOS) is produced at the source of transistor M2 1004. This results in a voltage between V_(ON) and 2V_(DD) being present at the gate of transistor M2 1004 through capacitance Cgs between the source and the gate of transistor M2 1004 (which capacitance is inherently present in transistor M2 1004), resulting in a voltage between ground and 4V_(DD) (represented by voltage 1012) being produced at the source of transistor M3 1006. This results in a voltage between V_(ON) and 4V_(DD) being present at the gate of transistor M3 1006 through capacitance Cgs between the source and the gate of transistor M3 1006, resulting in a voltage between ground and 6V_(DD) (represented by voltage 1014) being produced at the source of the transistor above transistor M3 1006. This process is repeated for the transistors going upward along the stack represented by transistors 1002, 1004, 1006, and 1008 until a voltage of between V_(ON) and 2(n−1)V_(DD) (represented by voltage 1016) is produced at the gate of transistor M_(n) 1008, which results in a voltage between ground and 2nV_(DD) being produced at the drain of transistor M_(n) 1008 and the output of the stack.

As stated above, whether the output of the stack is at ground or 2nV_(DD) depends on the input at the gate of transistor M1 1002. Referring back to FIG. 7, the corresponding inputs in the hybrid power DAC are at the gates of transistors M1 717. As shown, these input are controlled by the outputs of two-stack drivers 734, which are driven by inputs 702 and 704.

An example 800 of a two-stack driver that can be used for two-stack driver 734 in accordance with some embodiments is shown in FIG. 8. As shown, the input to the two-stack driver is provided at node 802 and the output is produced at node 804.

As shown in FIG. 7, the outputs of two-stack drivers 734 are connected to the gates of transistors M1 717 by match blocks 736. In some embodiments, match blocks 736 can include any suitable components for matching the impedance of the points on the left and right of each match block. For example, in some embodiments, match blocks 736 can include inductors, spirals, transmission lines, and/or capacitors.

As also shown in FIG. 7, the gates of transistors M1 717, M2 718, M3 720, and M4 722 are biased by bias voltages V_(g1), V_(g2), V_(g3), and V_(g4). These bias voltages can be produced in any suitable manner. For example, in some embodiments, for each hybrid power DAC 700, an adaptive bias circuit, such as adaptive bias circuit 900 shown in FIG. 9, can be provided.

As illustrated in FIG. 9, circuit 900 includes a voltage divider 902 formed from four resistors R₁. These resistors can have any suitable value. The voltage divider is powered by V_(DD,PA), which as described in FIG. 7 is variable and controlled by the output of inverters 710. Voltages V1, V2, and V3 shown in FIG. 9 are provided to transistors 904, 906, and 908, respectively. The bias voltages are then produced at the nodes labelled V_(g1), V_(g2), V_(g3), and V_(g4).

Resistors R_(big) can have any suitable values sufficiently large compared to the gate impedance (of the gates connected to the corresponding bias voltage) to have suitable performance but not too large so as to affect modulation speed. In some embodiments, R_(big) can be a 1 kΩ resistor.

In some embodiments, transistor 910 can be implemented as a bank of parallel binary weighted transistors (e.g., like the binary weighted transistors described above in connection with FIGS. 4 and 7) so that the bias voltages produced by circuit 900 can be controlled by controller 174 of FIG. 1. In some of these embodiments, any suitable number of parallel binary weighted transistors can be provided, and each of the transistors can have any suitable weighting(s). When such control is not needed, transistor 910 can be a single transistor.

Although specific components having specific properties (e.g., resistances, capacitance, sizes, relative sizes, voltages, etc.) are shown in FIGS. 1-10, one or more of the components in any one or more of these figures can be omitted or substituted with one or more alternate components having one or more different properties, in some embodiments.

The provision of the examples described herein (as well as clauses phrased as “such as,” “e.g.,” “including,” and the like) should not be interpreted as limiting the claimed subject matter to the specific examples; rather, the examples are intended to illustrate only some of many possible aspects.

Although the invention has been described and illustrated in the foregoing illustrative embodiments, it is understood that the present disclosure has been made only by way of example, and the numerous changes in the details of implementation of the invention can be made without departing from the spirit and scope of the invention, which is only limited by the claims which follow. Features of the disclosed embodiments can be combined and rearranged in various ways. 

What is claimed is:
 1. A circuit for a transmitter, comprising: a digital-to-amplitude converter (DAC) having an input and an output, comprising: a plurality of binary weighted switching transistors, each having a gate coupled to one of a plurality of amplitude control bits, a drain, and a source, and wherein the drain of each of the plurality of binary weighted switching transistors are connected together and wherein the source of each of the plurality of binary weighted switching transistors are connected together; a first transistor having a gate coupled to an input signal and a first bias voltage, a source coupled to the drains of the plurality of binary weighted switching transistors, and a drain; a second transistor having a gate coupled to a second bias voltage, a source coupled to the drain of the first transistor, and a drain; a third transistor having a gate coupled to a third bias voltage, a source coupled to the drain of the second transistor, and a drain; a fourth transistor having a gate coupled to a fourth bias voltage, a source coupled to the drain of the third transistor, and a drain; and an inverter having an input coupled to another amplitude control bit and having an output coupled to the output of the DAC and the drain of the fourth transistor; and an antenna coupled to the output of the DAC. 